This audio power amplifier in principle is a Current Feedback design (opposite to most amplifiers that are Voltage Feedback designs), see figure 1 below. The input signal is fed to an input buffer with a voltage gain of one and output impedance R0.
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The resistors RF and RI attenuate the output signal and make the feedback network. The difference between the input signal and the feedback signal results in the error current I. This is led to a current mirror and transformed to an equivalent voltage across the resistor RT, which normally has a high value. This voltage is fed to the output via an output buffer, which is a current amplifier (with a voltage gain of one). CT is a compensation capacitor, which provide for the stability of the amplifier (and the necessary phase margin). Amplifiers of the Current Feedback design are taken out to get amplifiers where the bandwidth is nearly independent of the voltage gain. Amplifiers of this type have a very large bandwidth and are very fast. |
The amplifier, view the circuit diagram, is fully symmetric built up as a voltage amplifier followed by a current amplifier. The relation R14/R15 approximately gives the closed loop gain.
The input stage consists of two JFETs in a symmetrical coupling instead of the usual (minimum four) bipolar transistors. If one extract the difference between the drain current for these complementary field effect transistors, in theory a non-distortion amplifier can be realised. In practice the two transistors will be different, causing the difference current to superimpose even harmonic components. As it is used bipolar transistors in a symmetric coupling to make the difference current, these will add additional odd harmonic components. The final result is an amplifier reminding of a valve amplifier: The distortion consists of mostly second harmonic components followed by third harmonic etc, in other words symmetric falling harmonic components.
The offset adjustment is made up by the potentiometer P13. The resistor R0 set the input impedance. The resistor R1 will together with the preamplifiers output resistance, the capacitor C1 and the input capacitance of the amplifier set the upper cut-off frequency. If the preamplifier has e.g. 600 ohms output impedance, the cut-off frequency is about 600 kHz.
The field effect transistors withstand a maximum of 20 V to operate properly and their internal capacitances are high. Using the cascode coupling as shown solves these problems. The common base transistors Q9/Q10 are bipolar types. It is also used 'base degeneration' by means of the resistors R3-6 instead of using a constant voltage on the bases of the transistors. The distortion from these transistors is very low compared to the relative high (even order) distortion from the field effect transistors.
Instead of using an ordinary current mirror for the current from Q9/Q10, here is used an amplifying current mirror. The voltage on the input of the amplifier is transformed to a proportional current in the JFETs. This current is compared to the feedback current (via R14). The resulting error current is transformed to a voltage across R7 (and R8). This error voltage is found across R19 (and R20) with little degradation, since Q16 (and Q17) operates with constant current (and with that constant base-emitter voltage). The error current is amplified as the relation given by R7/R19 (and R8/R20), and is transformed to a proportional voltage in the summing point before the current amplifier.
The open loop gain is given by the JFET's transconductance reduced with local feedback, multiplied with the relation R7/R19 and finally multiplied with the double load on the base of Q33/Q34. Without the resistors R23 and R24 the load at this point is both load dependent and parameter dependent. Thus these resistors determine the open loop gain to about 46 dB. With a closed loop gain at about 26 dB, the feedback factor is relatively low: about 20 dB.
Instead of placing the compensation capacitor to ground, these (two are used because of the symmetry) are placed between the input and output of the last voltage gain stage. Due to the Miller effect it is possible to use small valued capacitors (C25 and C26), which in turn means higher Slew Rate value. In this case the Slew Rate limitation do not set in before over 100 V/µs (without input and output filter).
The amplifier has the same open loop gain and bandwidth for all audio frequencies, since the open loop bandwidth is very high, nearly 100 kHz. This is equivalent with nearly equal amount of distortion and the same output impedance over the whole audible range. The components 23-26 is chosen to give a phase margin of 90 degrees at the given closed loop gain. The linearity is good, thanks to the use of an “inverted” Compound-coupling (Q16/Q21 and Q17/Q22). The distortion is less than it would be if an ordinary current mirror were used (without an additional buffer).
The current amplifier used is a modified Compound emitter follower, chosen from its good linearity and thermal stability. The bias generator consists of the components 27-30. The quiescent current is set by means of the potentiometer P29. The components 43-45 realise a Thiele LRC network to reduce the problem with high frequency instability when the load is capacitive and at the same time is reducing radio frequency induction into the loudspeaker connection. The output transistors in principle look into a constant resistive load and are protected against the most serious fluctuations of the high frequency variation of the loudspeaker impedance. The nominal load impedance is 8 ohms, and the cut off frequency of the network is approximately 500 kHz.
The unregulated power supply to the voltage amplifier is low pass filtered by means of R50 and R51 plus C48 and C49. Together with the decoupling capacitors C46 and C47 and the feedback, the low pass filtering ensures that ripple and noise on the supply voltage not reach the amplifier output.
The used power supply for the current amplifier is common for the two channels. It is however used separate supply for the current and voltage amplifier, view the schematic. Since this is a class A amplifier with global feedback, a common supply is sufficient when the filtering capacitors are large enough. These are Computer Grade type to ensure long lifetime.
The power supply voltage for the voltage amplifier is about 5 V higher than for the current amplifier. Higher power output is thus achieved without higher power dissipation worth mentioning (for an equal value of the power supply voltage to the current amplifier). Using the values in the parts list, the amplifier will perform 30 W RMS into 8 ohms. This demands a quiescent current of 1.4 A. The quiescent current may be increased above this value with the used heat sinks. In the prototype this value is set to 1.6 A, this corresponds to class A working for full output voltage down to a load impedance of 6 ohms.
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Some measuring results |
The layout and component placement are shown as images. The printed circuit board measures 139x76 mm. Two boards are needed for a stereo version. The parts list applies for both the circuit board and the power supply. |
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Output power: |
2x30 W RMS |
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The coil L46 can be made of 1 mm sealed copper wire. The diameter should be 12.5 mm, and the number of windings should be 20. With the exception of the power resistors, in the prototype it was used 1/2 W metal film oxide resistors with 1 % tolerance. If one wishes to use the far more expensive Holco (or even more expensive Vishay) types, it is recommended to first try the resistors 0, 1, 3-6, 14 and 15. In the prototype the JFETs 2SK170GR/2SJ74GR are used for J11/J12. The couple 2SK147GR/2SJ72GR may be used instead. These have higher internal capacitances and transconductance and are slightly more linear. In return the price is higher. For both JFETs one may use the BL-types, which have higher saturation current, but otherwise are equivalent. A replacement for the couple 2SC1775/2SA872 used for Q9/Q10 is e.g. 2SC1815/2SA1015 or 2SC2240/2SA970. The transistors Q16/Q17 may also be replaced by 2SA970/2SC2240. 2SC2238/2SA968 is used for Q21 and Q22. These may be replaced by 2SC4793/2SA1837. For all replacements, be sure that the pinning is correct when mounting the transistors on the circuit board.
In the prototype is Q21 and Q22 (2SC2238/2SA968) mounted with a small heat sink. The bias generator is also 2SC2238 (Q30). The output transistors used for Q41/Q42, are the well-known couple 2SA1216/2SC2922 from Sanken. The author has not been watching for any replacement for these. They are relatively linear, fast and at the same time very rugged, a seldom combination. These transistors have been on the market for quite a long time now and are relatively often used in commercial amplifiers, and they are relatively cheap. They come in a rare plastic housing, something that makes it necessary to mount them directly to the heat sink. In the prototype the printed board is fastened to the heat sink, and the transistor leads are fastened to the board on the solder side. This is also done for the drivers (Q33/Q34) and the bias transistor (Q30). A better solution would be to have separate cooling for the drivers, but the quiescent point is not drifting much when the temperature has stabilised. Besides additional heat sinks are spared in this way.
For the power supply it is used a common copper plate as ground plane. One terminal of each electrolytic capacitor is screwed on this plate (Remember the polarity!). The transformer mid point (normally it is two conductors) is also screwed on this plate. From the plate it is necessary to have a good connection to chassis. The phono socket ground terminal is connected to chassis (near the input). The (ground) shield of the phono cable is connected to the circuit board point marked with 'SG' (Signal Ground). The inner conductor of the phono cable is connected to the circuit board point marked with 'IN'. The loudspeaker output minus socket is connected to the chassis at the output. From the loudspeaker output the two conductors are twisted and fastened to the circuit board in the two points marked with 'OUT' and 'PG' (Power Ground). The last is connected to the minus conductor. All connections should be as short as possible. If some sort of instability or noise should occur, the probability is high that the reason is bad wiring (e.g. earth loops).
It is recommended to use a variable transformer or variable DC voltage generator first time the amplifier is started up. When the power supply voltage is increased, adjust the output-offset voltage by means of the potentiometer P13 to be close to 0 V DC. Also adjust the quiescent current to initially be at a minimum, and increase this slowly by means of the potentiometer P29. If possible, look at the output with an oscilloscope, there should not be anything but noise here if everything is OK. When the temperature is increasing, it is necessary to re-adjust both offset voltage and quiescent current (min. 1.40 A). The offset voltage at the output varies, but should not exceed 30 mV.
The amplifier is not provided with any servo coupling. If one has loudspeakers which not at all can withstand any offset voltage (rarely a problem), one may add a bipolar (or bipolar coupled) electrolytic in series with R6. The value would be very high, for a cut off frequency of 5 Hz the value is 330 µF. It would be a good idea to add a high quality plastic capacitor in parallel (e.g. with a value of 100 nF). It is not made any room to these capacitors on the circuit board.
For the 30 W prototype about 0.75 V RMS input voltage is required for full output power. This should be sufficient for the most modern signal sources without being forced to use a preamplifier. If higher gain is wanted, R15 is reduced (and vice versa). Please note that the feedback resistor R14 should be unchanged. None of the good properties of the amplifier, like bandwidth, distortion and Slew Rate, are deteriorated by moderate change of R15.
The author has not performed any experiments with class A amplifiers with higher output power than 50 W RMS. The cooling requirement is large and should not be underestimated. It is the author’s belief that 25 W RMS real class A is sufficient for domestic use in most cases. A class A amplifier generally is perceived more powerful than a class B (or A/B).
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Copyright©2001 Knut Harald Nygaard |